Method and device for downconverting the sampling frequency of a digital signal, for example in a non-integer frequency ratio

ABSTRACT

A method for converting a sampling frequency of a digital signal sampled at a first sampling frequency includes receiving digital signal input samples, and forming output samples corresponding to a second sampling frequency based on the digital signal input samples and an interpolation filter. The first sampling frequency may be larger than the second sampling frequency. The method may further include delivering the output samples. Forming output samples includes, for each of the digital signal input samples, updating current values of N successive output samples with N contributions. The N contributions may be respectively calculated based on a value of a current input sample of the digital input samples weighted by values of N filter coefficients associated with the current input sample, N being fixed and identical for all the digital signal input samples regardless of a value of the conversion ratio between the first and second sampling frequencies.

FIELD OF THE INVENTION

The invention relates in particular to the down conversion of thesampling frequency of a digital signal, also called “sub-sampling” or“decimation”, in a, for example, non-integer, frequency conversion ratiobetween the input sampling frequency (which is the higher) and theoutput sampling frequency (which is lower).

The invention applies advantageously but non-limitingly to wirelesscommunication systems and, more particularly, to multi-standardreception platforms which use multiple different sampling frequencies.

BACKGROUND OF THE INVENTION

The WiMAX standards (acronym for “Worldwide Interoperability forMicrowave Access”), which is a family of standards defining high-speedRF connections intended mainly for point-multipoint architectures, maybe cited in this regard. The WiMAX standards comprise in particular thestandards of the 802.16 family.

The standards of the 802.11 family, also known by the name “WiFistandards”, may also be cited. The IEEE 802.11 standard is aninternational standard describing the characteristics of a wirelesslocal area network (WLAN) capable of communicating at high speed over aradius of several tens of meters indoors.

The production of a receiver capable of being compatible with severaltransmission standards generally requires the ability to processmultiple sampling frequencies of the digital signal. Because of thefrequency synthesis constraints, the sampling frequency of the signal atthe level of the analog/digital conversion stage is not necessarilyflexible.

Moreover, analog/digital converters of low power, that is to say of lowcurrent consumption, make increasing use of oversampling to reduce thenumber of quantization levels while maintaining a goodsignal/quantization noise ratio. This type of analog/digital convertertherefore generally requires that the quantization noise be filteredoutside of the useful band of the signal and that the sampling frequencybe reduced to a lower value than that at which the digital signal isprocessed. Therefore, conventional sampling frequency conversion devicesgenerally require a certain number of multiplications at high frequency,which leads to considerable complexity of the device and consequently tosignificant power consumption.

Moreover, it is desirable to be able to perform a sampling frequencyconversion in a non-integer conversion ratio, so as to be capable ofproviding a sampling frequency that is compatible both with the desiredtransmission standard and with the sampling frequency of theanalog/digital conversion stage.

Several approaches exist for performing a sampling frequency conversionwith a non-integer conversion ratio.

A first approach includes converting the digital signal into an analogsignal, using a digital/analog converter, then filtering the analogsignal, and thereafter reconverting it into a digital signal with thedesired sampling frequency. Such an approach is hardly satisfactorysince it leads to significant conversion noise and there is hardly asatisfactory method for generating the clock signal of theanalog/digital converter allowing the reconversion of the analog signalto the desired sampling frequency.

Another approach includes oversampling the digital signal by a factor S,in filtering at the same time the signal aliasing components and thequantization noise, and in thereafter performing a decimation (samplingfrequency down conversion), by a factor M. A sampling frequencyconversion in a rational ratio S/M is then obtained. However, such anapproach is not truly flexible. Indeed, only a small number ofconversion ratios can be foreseen, since each conversion ratio mayrequire the production of a filter. Furthermore, generally only rationalfrequency conversion ratios (ratios whose numerator and denominator areintegers) can be used. Furthermore, with operating frequencies currentlyalready being very high, it is particularly difficult to carry out anoversampling without increasing the power consumption and the overallcomplexity of the device.

Another approach resides in an interpolation, or in a calculation of newsamples (interpolated samples) on the basis of the input samples byusing an interpolation filter. However, such approaches are currentlycomplicated and use significant memory resources, as well as high-speedbuses, for storing all the coefficients of the filter as a function ofthe temporal deviation between an interpolated sample and the inputsamples, which serve to calculate this interpolated sample.

Moreover, in certain applications, stages of the delta-sigma type areadvantageously used for the analog/digital conversion stages forgenerating the digital signal whose sampling frequency is to beconverted. Such analog/digital converters of delta-sigma type areparticularly beneficial since they make it possible to code the digitalsignal on a low number of bits while rejecting the quantization noiseoutside of the useful band of the signal. The current approaches forsampling frequency down conversion (decimation) are not suited tooversampled signals originating from converters of the delta-sigma typesince the current approaches generally perform only a filtering of thesignal aliasing components, but not of the quantization noise.

SUMMARY OF THE INVENTION

According to one mode of implementation and embodiment, a method and adevice for converting a sampling frequency of a digital signal isproposed. The device and method make it possible to perform a frequencydown conversion in a rational or irrational, integer or non-integerratio, which is adjustable, thereby making it possible to readily adaptto transmission multi-standard reception platforms.

There is also proposed, according to one mode of implementation andembodiment, a method and device for converting a sampling frequencyusing an interpolation filter with a low complexity of production andimplementation.

There is also proposed, according to another mode of implementation andembodiment, a method and device for converting a sampling frequencywhich allows filtering of the replicas of the signal and also of anycomponent of the signal which might be aliased in the useful band of thesignal after the sampling frequency conversion. Thus, it is possible tofilter the quantization noise arising, for example, from ananalog/digital conversion stage.

There is also proposed, according to another mode of implementation andembodiment, a method and device for converting a sampling frequency inwhich no approximation on the form of the filter is performed. Thus itis possible to take exact filtering coefficients into account.

According to one aspect, a method for converting a sampling frequency ofa digital signal sampled at a first sampling frequency includes:reception of input samples of the digital signal, and formulation ofoutput samples corresponding to a second sampling frequency on the basisof the input samples and of an interpolation filter. The first samplingfrequency is larger than the second sampling frequency. The method alsoincludes delivery of the output samples.

According to this aspect, the formulation of the output samples includesfor each current input sample, the updating of the current values of Nsuccessive output samples with N contributions, respectively calculatedon the basis of the value of this current input sample, and weighted bythe values of N filter coefficients associated with this current inputsample. This number N is fixed and identical for the input samples, nomatter the value of the conversion ratio between the two frequencies,which can be arbitrary, for example, integer or non-integer.

Stated otherwise, according to this approach, and unlike the existingapproach in which the impulse response of the filter is based on a firsttime period corresponding to the first sampling frequency (therebyimplying that an output sample should be calculated with a fixed numberof input samples), the interpolation filter is based on the second timeperiod corresponding to the second sampling frequency, therebysignifying that an input sample contributes to the determination of thevalues of a fixed number of output samples. This fixed number is alsoindependent of the conversion ratio. This is not the case for theexisting approaches, which consequently involve storage of filtercoefficients for each conversion ratio.

On the other hand, if each current input sample contributes to thedetermination of the values of a fixed number of output samples, anoutput sample can receive a different number of contributions from thenumber of contributions intended for another output sample.

Thus, it is possible to filter not only the replicas of the sampledsignal, but also any component which might be aliased in the useful bandafter conversion of the sampling frequency.

Even if the value of the conversion ratio can be arbitrary, there is aparticularly beneficial mode of implementation for non-integer, rationalor irrational values of the conversion ratio, since this affords inparticular an approach and a hardware embodiment, even for non-integerconversion ratios.

Moreover, the coefficients of the filter can be, for example,advantageously calculated “on line.” In other words, the coefficientscan be calculated when they are needed for the calculation of thecontributions.

According to one mode of implementation, an output sample is delivered,whose value has been updated with a contribution originating from acurrent input sample, if this output sample should not receive anycontribution originating from the following input sample. The effectivevalue of this delivered output sample is then the value updated with thecontribution originating from the current input sample.

According to one mode of implementation, the width of the impulseresponse of the interpolation filter fixes the value of the number N.Thus, for example, an interpolation filter is chosen, the width of whoseimpulse response is equal to the product of a multiplying coefficienttimes the second time period corresponding to the second samplingfrequency, and the value of N is fixed at a value equal to the higherinteger part of the multiplying coefficient.

According to an exemplary implementation, during the updating of thevalues of the output samples of ranks n to n+N−1, these N output samplesare respectively assigned N storage of ranks r₁ to r_(N) in each ofwhich is accumulated, for the output sample considered, the variouscontributions arising from the various input samples. The delivery ofthe output sample of rank n with its effective value includes thedelivery of the value included in the storage of rank r₁. The assignmentof the registers of ranks r₁ to r_(N−1) is to the output samples ofranks n+1 to n+N−1, the assignment of the register of rank r_(N) is tothe output sample of rank n+N, and the initialization of the register ofrank r_(N) to a chosen initialization value, for example the zero value.An exemplary implementation, such as this, is particularly simple tocarry out.

According to one mode of implementation, the input samples are receivedin time with the edges of a clock signal having a frequency equal to thefirst sampling frequency, and the output samples are formulated in timewith the edges of this clock signal. On the other hand, the outputsamples are delivered on only some of the edges of the clock signal.This depends in particular on the number of contributions forformulating the effective value of an output sample. Thus, the outputsamples delivery frequency can be different from the second samplingfrequency. These delivered output samples actually correspond to asampling of the digital signal at the second sampling frequency.

The value of the sampling frequency conversion ratio is advantageouslyselected from among a suite of values corresponding to values of secondsampling frequency that are compatible with several different standards,for example the WiMAX or WiFi standards.

An interpolation filter is also advantageously chosen whoseamplitude/frequency response curve exhibits zeros at the frequencieswhich are multiples of the second sampling frequency.

This makes it possible to have good attenuation in frequency bandssituated around the multiples of the second sampling frequency. This isparticularly beneficial when the analog/digital conversion of the signalis of the delta-sigma type. Indeed, such a filter makes it possible tofilter the quantization noise effectively.

It is, for example, possible to use an interpolation filter whoseamplitude/frequency response curve exhibits a form of the type sinc2·cos (sin c2 denotes the cardinal sine function squared). This makesit possible to obtain an attenuation of at least 60 dB in the bandsmentioned above.

According to one mode of implementation, the calculation of the Ncontributions of a current input sample includes the determination ofthe temporal position of this current input sample with respect to the Ntemporal instants, termed “temporal instants of interpolation”, spacedmutually apart by the second time period corresponding to the secondsampling frequency and respectively associated with the N updated outputsamples, and the consideration of the temporal position in thedetermination of the values of the N coefficients of the filter.

More precisely and by way of example, if the input samples arrive atfirst temporal instants, the determination of the temporal position of acurrent input sample with respect to the N temporal instants ofinterpolation comprises the determination of a temporal interval betweenthe first temporal instant associated with this current input sample andone of the N temporal instants of interpolation, and the determinationof the other temporal intervals between the first instants and the N−1other temporal instants of interpolation on the basis of the temporalinterval, and the values of the N coefficients of the filter correspondrespectively to the values of the impulse response of the filter atthese N temporal intervals.

Moreover, the fact of basing the impulse response of the interpolationfilter on the value of the second time period, and the fact of usingthis second time period as a temporal reference base for theinterpolation, enables the determination of the values of thecoefficients of the filter on the basis of a function of the temporalinterval.

This function can comprise, for example, at least one linear zone, thisbeing the case when the filter has an amplitude/frequency response curveexhibiting a form of the type sin c2·cos. This then allows a significantreduction in production complexity.

Actually, according to one mode of implementation, the value of thetemporal interval varies between 0 and 1 and an auxiliary temporalinterval is defined on the basis of this temporal interval, whose valueis equal to that of the temporal interval referred to the span [0;0.25], and the calculation of a contribution of a current input samplecomprises a single multiplication of the value of this input sample withthe value of the auxiliary temporal interval.

According to another aspect, there is also proposed a method forprocessing an analog signal, comprising an analog/digital conversion ofthe analog signal and a method for converting sampling frequency asdefined previously, of a digital signal arising from the analog/digitalconversion. The analog/digital conversion can be of the delta-sigmatype.

Moreover, the analog signal can be a signal chosen from among severalsignals respectively compatible with several transmission standards, forexample the WiMAX or WiFi standard.

According to another aspect, there is also proposed a device forconverting the sampling frequency of a digital signal sampled at a firstsampling frequency.

This device comprises a receiver able to receive input samples of thedigital signal, a formulator able to formulate the output samplescorresponding to a second sampling frequency on the basis of the inputsamples and of an interpolation filter, the first sampling frequencybeing larger than the second sampling frequency, and an output able todeliver output samples. The formulator comprising a calculator able foreach current input sample to calculate N contributions respectively onthe basis of the value of this current input sample weighted by thevalues of N filter coefficients associated with this input sample, thisnumber N being fixed and identical for all the input samples whateverthe value of the conversion ratio between the two frequencies, andprocessor able for each input sample to update the current values of Nsuccessive output samples with the N contributions.

According to one embodiment, the formulator further comprises acontroller to authorize the delivery of an output sample whose value hasbeen updated with a contribution originating from a current input sampleif this output sample should not receive any contribution originatingfrom the following input sample, the effective value of this deliveredoutput sample being the value updated with the contribution originatingfrom the current input sample.

According to one embodiment, the processor comprises N inputs able toreceive respectively the N contributions of a current input sample, Nadders each possessing a first input coupled to the corresponding inputof the processor and a second input, N multiplexers able to becontrolled by the controller and each possessing a first input connectedto the output of the corresponding adder and a second input, a chain ofN registers each possessing an input coupled to the output of thecorresponding multiplexer and an output coupled to the second input ofthe corresponding adder, the output of the last register of the chainbeing coupled to the output means and the output of each of the otherregisters being coupled to the second input of the multiplexerassociated with the following register in the chain, an auxiliaryregister containing a zero value data item and coupled to the secondinput of the multiplexer associated with the first register of thechain.

According to one embodiment, the receiver is able to receive the inputsamples in time with the edges of a clock signal having a frequencyequal to the first sampling frequency and the formulation means are ableto formulate the output samples in time with the edges of the clocksignal and the output means are able to deliver the output samples ononly some of the edges of the clock signal.

According to one embodiment, the device can furthermore comprise aselector able to select the value of the ratio from among a suite ofvalues corresponding to values of second sampling frequencies that arecompatible with several different standards, for example the WiMax orWiFi standards.

According to one embodiment, the calculator comprises a first detectorable to determine the temporal position of a current input sample withrespect to the N second instants, termed temporal instants ofinterpolation, respectively associated with the N output samplesintended to be updated, and a second detector able to determine thevalues of the N coefficients of the filter on the basis of the temporalposition.

According to one embodiment, the first detector is able to determine atemporal interval between the first instant associated with a currentinput sample and one of the N temporal instants of interpolation, and todetermine the other temporal intervals between the first instant and theN−1 other temporal instants of interpolation on the basis of thetemporal interval, and the values of the N coefficients of the filtercorrespond respectively to the values of the impulse response of thefilter at these N temporal intervals.

The second detector may be able to determine each of the N coefficientsof the filter on the basis of a function of the temporal interval, thisfunction possibly being, for example, a linear function.

In this case, the value of the temporal interval varying between 0 and1, the first detector is able to determine an auxiliary temporalinterval on the basis of this temporal interval whose value is equal tothat of the temporal interval referred to the span [0; 0.25], and thecalculator comprises a single multiplier possessing an input coupled tothe input means and a second input coupled to the first detector forreceiving the auxiliary temporal interval.

According to one embodiment, the calculator further comprises a blockcomprising at least one an adder, at least one subtractor and at leastone shifter for shifting bits that are individually selectable as afunction of the value of the temporal interval.

According to another aspect, there is also proposed a device forprocessing an analog signal comprising an input for receiving the analogsignal, an analog/digital converter coupled to the input, and a samplingfrequency conversion device such as defined above, whose input means arecoupled to the output of the analog/digital converter. Theanalog/digital converter can be of the delta-sigma type.

According to another aspect, there is also proposed a receiver belongingto at least one wireless communication system, comprising an antenna forreceiving an incident signal and a processing device such as definedabove.

The receiver can thus form a multi-standard reception platform capablefor example of processing signals compatible with the WiMAX or WiFistandard.

Other advantages and characteristics of the invention will becomeapparent on examining the detailed description of wholly non-limitingmodes of implementation and embodiment and the appended drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 schematically illustrates an embodiment of a multi-standardreceiver incorporating an exemplary sampling frequency conversion deviceaccording to the invention.

FIGS. 2 and 3 schematically illustrate exemplary implementations of amethod for converting sampling frequency according to the invention.

FIGS. 4 and 5 illustrate an exemplary implementation of the inventionwith an exemplary interpolation filter according to the invention.

FIG. 6 schematically illustrates an exemplary determination of thecontributions of an input sample according to the invention.

FIGS. 7 to 9 more precisely illustrate an example of characteristics ofan interpolation filter according to the invention.

FIG. 10 illustrates in greater detail an exemplary embodiment of asampling frequency conversion device according to the invention.

FIGS. 11 to 13 illustrate an exemplary manner of operation of the deviceof FIG. 10.

FIGS. 14 and 15 illustrate curves of evolution of power consumed.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

In FIG. 1, the reference REC denotes a multi-standard receiver capableof processing signals compatible with various transmission standards, asfor example the WiMAX and WiFi standards.

More precisely, this reception platform REC comprises an antenna ANTcapable of receiving an incident signal compatible with one of thetransmission standards. An analog processing stage ETA is connected tothe antenna ANT and comprises, in particular, mixers capable ofperforming a frequency transposition so as to return, for example, theanalog signal to baseband according to two quadrature processingpathways, namely the “I” (in phase) pathway and the “Q” (phasequadrature) pathway according to a conventional denomination known tothe person skilled in the art.

To this analog processing stage ETA is coupled an analog/digitalconversion stage ADC, preferably of the delta-sigma type. The structureof such an analog/digital conversion stage of the delta-sigma type isconventional and well known to the person skilled in the art.

A digital processing stage ETN is coupled to the output of theanalog/digital conversion stage ADC and comprises in particular asampling frequency conversion device DCF followed by other processingmodules, for example an FFT module, embodied, for example, within aprocessor, commonly called a “baseband processor” by the person skilledin the art.

In wireless communication systems using, for example, a modulation ofthe OFDM type Orthogonal Frequency Division Multiplexing, which is amodulation of digital signals by orthogonal frequency division, thesampling frequency of the analog digital converter is not necessarily amultiple of the sampling frequency for the FFT block and specified bythe standards.

By way of indication, for the WiFi-WiMAX standards, the samplingfrequency of the signal fed to the FFT module can be chosen from amongthe following values for the WiFi standards: 10-20 MHz, for thestandards WiMax: 4; 5.6; 8; 10; 11.2; 22.4 MHz.

It is therefore apparent in this application that the sampling frequencyconversion device DCF should be capable of performing a frequencydecimation, that is to say a sampling frequency down conversion, in anon-integer ratio, and not a rational ratio. This conversion ratio maytherefore be a non-integer real number greater than 1, although integernumbers greater than 1 are also possible.

Furthermore, as seen in greater detail below, the device DCF is capableof selecting the conversion ratio F_(s)/F_(i) from a set ofpredetermined values, for example, the set of values mentioned above.The frequency F_(s) denotes the sampling frequency of the signalupstream of the device DCF, and is also subsequently called the “firstsampling frequency”, while F_(i) denotes the sampling frequency of thedigital signal delivered by the device DCF, also called the “secondsampling frequency”.

Subsequently in the text, by way of example, that the digital signalx(kT_(s)) received by the device DCF, is a baseband signal, having alimited frequency band (varying from −B/2 to +B/2 where B denotes thebandwidth of the signal) greatly oversampled with a sampling frequencyF_(s)=1/T_(s). This sampling frequency F_(s) is very much greater thanB/2, for example of the order of ten to fifteen times B/2.

y_(i)(nT_(i)) denotes the digital signal sampled at the frequency F_(i)after conversion of the sampling frequency.

As seen in greater detail below, this sampling frequency conversion isperformed on the basis, in particular, of an interpolation filter. Inthis regard, h_(I)(t) denotes the impulse response of the interpolationfilter in the time domain.

The frequency spectrum X(f) of the input signal is periodic with aperiod F_(s), thereby signifying that there are replicas of the signalat frequencies spaced apart by F_(s). Moreover, the filtering operationis equivalent to a discrete sum.

The output digital signal, resampled at the frequency F_(i), can beexpressed by formula (I) below:

$\begin{matrix}{{y_{i}\left( {nT}_{i} \right)} = {{y(t)} \cdot {\sum\limits_{n}{\delta\left( {t - {nT}_{i}} \right)}}}} & (I)\end{matrix}$

The equivalent of this formula (I) in the frequency domain is expressedby the formula (II) below:

$\begin{matrix}{{Y_{i}(f)} = {\left\lbrack {{X(f)}{{xH}_{I}(f)}} \right\rbrack*{\sum\limits_{n}{\delta\left( {f - {nF}_{i}} \right)}}}} & ({II})\end{matrix}$

This equation shows the well known effect of aliasing. Each signalcontained in a band of frequencies [rF_(i)−B/2; rF_(i)+B/2] are added,because of the convolution, to the useful signal contained in thefrequency band [−B/2; +B/2].

Subsequently in the text, for the sake of simplification, x_(k) denotesthe samples of the input signal and y_(n) denotes the output samples.

Referring now more particularly to FIG. 2, it is seen than the inputsamples are received (step 20) at the first sampling frequency F_(s).

The output samples y_(n) corresponding to the second sampling frequencyF_(i) are formulated on the basis of the input samples and of aninterpolation filter whose impulse response in the time domain is equalto h_(I).

The ratio F_(s)/F_(i) is strictly greater than 1 and is, for example, arational or irrational non-integer number.

The formulation of the output samples includes for each current inputsample x_(k), the updating (step 22) of the current values of Nsuccessive output samples with N contributions respectively calculated(step 21) on the basis of the value of the current input sample x_(k)weighted by the values of N filter coefficients h1, . . . hN, which areassociated with this current input sample. It should be noted that thenumber N is fixed and identical for all the input samples x_(k),whatever the value of the ratio F_(s)/F_(i).

Moreover, as seen in greater detail below, the delivery 23 is performedof an output sample y_(n) whose value has been updated with acontribution originating from a current input sample x_(k) if thisoutput sample should not receive any contribution originating from thefollowing input sample x_(k+1). In this case, the effective value ofthis delivered output sample is the value updated with the contributionoriginating from the current input sample x_(k).

In the converse case, that is to say if the sample y_(n) should receivea contribution originating from the following input sample x_(k+1),steps 21 and 22 are repeated for the sample x_(k+1) (step 24).

This is illustrated more precisely in FIG. 3. The contribution for theupdating 31 of the sample y_(n) is equal to the product 30 of the valueof the current input sample x_(k) times the value h_(j) of thecorresponding coefficient of the filter.

After updating the sample y_(n) with this contribution, it is determined(step 32) whether or not the following input sample x_(k+1) should makea contribution to the value of the output sample y_(n).

If the response is no, then the output sample y_(n) is delivered withits effective value equal to its last updated value. If the response isyes, the new contribution is calculated for the following input samplex_(k+1) (step 34) in a manner analogous to that which has just beendescribed for the current input sample x_(k).

In a general manner, the width of the impulse response of theinterpolation filter fixes the value of the number N. Thus, for example,if an interpolation filter is chosen, the width of whose impulseresponse is equal to the product cm·T_(i) (where cm denotes amultiplying coefficient, not necessarily integer, and T_(i) the secondtime period corresponding to the second sampling frequency F_(i)), thevalue N is equal to the higher integer part of the multiplyingcoefficient cm (N=┌cm┐).

In the example illustrated in FIGS. 4 and 5, the interpolation filterhas a bandwidth equal to 2.5 T_(i). The number N is consequently equalto 3. In other words, each input sample contributes to the updating ofthree output samples.

It is also assumed in this example that the impulse response h_(I)(t) isan even function h_(I)(t) is equal to h_(I)(−t), and h_(I)(t) is equalto zero if t does not belong to the interval [−3T_(i)/2;3T_(i)/2].

The calculation of the N contributions (N=3 here) of each current inputsample, comprises the determination of the temporal position of thisinput sample x_(k) with respect to the N temporal instants, t_(n),t_(n+1), t_(n+2) (FIGS. 4 and 5) that is dubbed “temporal instants ofinterpolations”, which are spaced mutually apart by the second timeperiod T_(i) and are respectively associated with the N=3 updated outputsamples y_(n), y_(n+1), y_(n+2). Also, this temporal position is takeninto account in the determination of the values of the N coefficients ofthe filter (N=3 here).

This is illustrated more precisely in FIG. 6, in the general case N.More precisely, according to this implementation, and if the temporalinstants at which the input samples x_(k) arrive are termed “firsttemporal instants”, the determination of the temporal position of acurrent input sample x_(k) with respect to the N temporal instants ofinterpolation t_(n), . . . , t_(n+N−1), includes the determination of atemporal interval μ_(k) between the first temporal instant associatedwith the current input sample x_(k) and one of the N temporal instantsof interpolation, for example the previous temporal instant ofinterpolation.

In the case of FIG. 4, the previous temporal instant of interpolation isthe instant t_(n+1), while it is the instant t_(n) in the configurationillustrated in FIG. 5.

As illustrated in FIG. 6, the values of the N filter coefficients h1, .. . hN, with which the N contributions of the current input sample x_(k)is calculated, is determined (step 61) on the basis of the determination60 of μ_(k).

In the example illustrated in FIGS. 4 and 5, the calculation of thecoefficients h1, h2 and h3 depends on the position of μ_(k) with respectto the value 0.5. More precisely, if the value of μ_(k) is less than0.5, the configuration of FIG. 4 applies. In this case, it is seen thatthe three coefficients h1, h2 and h3 are respectively determined by thefollowing formulae (III) (IV) and (V):h1=h _(I)(1+μ_(k))  (III)h2=h _(I)(μ_(k))  (IV)h3=h _(I)(1−μ_(k))  (V)

It is noted in this regard that (1+μ_(k)) and (1−μ_(k)) represent twoother time intervals between the instant of arrival of the currentsample x_(k), and the other temporal instants of interpolation t_(n) andt_(n+2).

In the case where μ_(k) is greater than 0.5, the configurationillustrated in FIG. 5 applies. In this case, the coefficients h1, h2 andh3 are respectively given by formulae (VI), (VII) and (VIII) below:h1=h _(I)(μ_(k))  (VI)h2=h _(I)(1−μ_(k))  (VII)h3=h _(I)(2−μ_(k))  (VIII)

An exemplary algorithm for formulating the value of the output samplesmay be as follows for a current input sample x_(k), step a): calculationof the value of μ_(k) through the formula:μ_(k)=[μ_(k−1) +R](mod 1)  (IX)(in this formula, R denotes the fractional decimation ratio T_(s)/T_(i)which is less than 1 and mod denotes the modulo function).

Once the value of μ_(k) has been determined, it is possible to calculatethe filter coefficients h1, h2, h3 associated with this current inputsample x_(k), in the following manner (step b):

a. if μ_(k)≦0.5i. h1=h _(I)(1+μ_(k)) h2=h _(I)(μ_(k))h3=h _(I)(1−μ_(k))

b. otherwisei. h1=h _(I)(μ_(k)) h2=h _(I)(1−μ_(k))h3=h _(I)(2−μ_(k))

It is then possible to update the values of the three output samplesy_(n), y_(n+1) and y_(n+2) in the following manner (step c):y _(n) =y _(n) +h1·x _(k)y _(n+1) =y _(n+1) +h2·x _(k)y _(n+2) =y _(n+2) +h3·x _(k)

A check is carried out thereafter (step d) to verify whether the outputsample y_(n) is ready to be delivered with its effective value, that isto say whether or not it ought to receive a contribution originatingfrom the following input sample x_(k+1).

In this regard, a check is carried out to verify whether the followingcondition CD1 is satisfied, that is to say whetheri. (μ_(k)≦0.5 and μ_(k) +R≧0.5) or μ_(k) +R>1.5  (CD1)

If this condition CD1 is satisfied, it is then certain that y_(n) doesnot receive any contribution from x_(k+1). In this case, y_(n) is readyto be delivered.

If such is the case, the output sample y_(n) is actually delivered asoutput and, as seen in greater detail, a shift is performed of theregisters to which the output samples are assigned and in which thevarious contributions arising from the various input samples areaccumulated for the output sample considered.

In the converse case when the condition CD1 is not satisfied, forexample, that is to say if the output sample is not ready to bedelivered because it will receive a contribution from the followinginput sample x_(k+1), steps a) to d), which have just been described,are repeated for the following sample x_(k+1).

In particular, when the analog/digital converter ADC is a converter ofthe delta-sigma type, it is particularly beneficial to choose aninterpolation filter whose amplitude/frequency response curve exhibits“0”s at frequencies that are multiples of the second sampling frequencyF_(i).

Moreover, for a particular example of the form of shaping the converterADC of delta-sigma type, no attenuation of the signal is sought in theuseful band of the signal and at least 60 dB of attenuation is sought inthe bands [rF_(i)±B/2].

An acceptable approach then includes using a cardinal sine function (sinc) for the amplitude/frequency response curve of the filter. Severalapproaches are possible according to the attenuation level desired. Inthis regard, it is possible to use either a simple sin c function, theproduct of a sin c function and a cosine function, or a sin c2 function(cardinal sine squared), the product of a sin c2 function and a cosinefunction, a sin c3 function (cardinal sine cubed), the product of thissin c3 function and a cosine function, or other functions that do or donot combine the cardinal sine function.

By way of indication, to obtain at least 60 dB of attenuation in thedesired bands, it is possible to choose a form of the type sin c2·cos.

It is thus possible to choose the amplitude/frequency response curveH_(I)(f) defined by the formula below.

$\begin{matrix}{{H_{I}(f)} = {{\sin_{c}\left( \frac{f}{F_{i}} \right)}^{2} \cdot {\cos\left( {2\pi\frac{f}{4F_{i}}} \right)}}} & (X)\end{matrix}$which corresponds, after inverse Fourier transform, to an impulseresponse h_(i)(t) equal to the formula below.

$\begin{matrix}{{h_{i}(t)} = {{\Pi_{T_{i}}(t)}*{\Pi_{Ti}(t)}*{\frac{1}{2}\left\lbrack {{\delta\left( {t - \frac{T_{i}}{4}} \right)} + {\delta\left( {t + \frac{T_{i}}{4}} \right)}} \right\rbrack}}} & ({XI})\end{matrix}$

Such an amplitude/frequency response curve is illustrated in FIG. 7,while FIG. 8 represents the corresponding impulse response h_(I)(t).

As the impulse response of the interpolation filter is based on T_(i)and since T_(i) is also the base of the temporal reference for theinterpolation calculation (see FIGS. 4 and 5), it then becomes possibleto determine each of the N coefficients of the filter (N=3 in theexample) on the basis of a function of the temporal interval μ_(k).

This simplifies the calculation of the coefficients of the filter.

And further still, for a filter of the type of those using the productof a cardinal sine function and a cosinusoidal function, such as, forexample, that illustrated in FIGS. 7 and 8, this function can include atleast one linear zone.

This is illustrated in FIG. 9. In this figure, curve C1 represents theevolution of the coefficient h1 as a function of μ_(k). Curve C2represents the evolution of the coefficient h2 as a function of μ_(k),and curve C3 represents the evolution of the coefficient h3 as afunction of μ_(k). It is seen that these curves are formed of linearzones, in this instance, straight line segments.

It should also be noted that for the sake of clarification, the valuesof the coefficients h1, h2 and h3 have intentionally been multiplied by2 in FIG. 9.

In FIG. 9, it is seen that if the abscissa axis is subdivided into foursectors S1, S2, S3 and S4, as a function of the span of values of μ_(k),it is possible to define, as indicated in the table below, an auxiliarytemporal interval μ′_(k) whose value is equal to that of the temporalinterval μ_(k), but referred to the span [0; 0.25]. Stated otherwise,μ′_(k) always belongs to the span [0; 0.25].

TABLE 1 Span of μ_(k) μ_(k) ∈[0; 0.25] μ_(k) ∈[0.25; 0.5] μ_(k) ∈[0.5;0.75] μ_(k) ∈[0.75; 1] μ′_(k) = μ_(k) μ′_(k) = μ_(k) − 0.25 μ_(k)′ =μ_(k) − 0.5 μ_(k)′ = μ_(k) − 0.75 and R_(k) = 0 and R_(k) = 1 and R_(k)= 2 and R_(k) = 3 H₁ h₁(μ_(k)′) = 0.25 − μ′_(k) h₁(μ_(k)′) = 0h₁(μ_(k)′) = 1 − 2μ_(k)′ h₁(μ′_(k)) = 0.5 − μ′_(k) h₂ h₂(μ′_(k)) = 1.5h₂(μ′_(k)) = 1.5 − 2μ′_(k) h₂(μ′_(k)) = 0.5 + 2μ′_(k) h₂(μ′_(k)) = 1.5h₃ h₃(μ′_(k)) = 0.25 + μ′_(k) h₃(μ′_(k)) = 0.5 + 2μ′_(k) h₃(μ′_(k)) = 0h₃(μ′_(k)) = μ′_(k)

In the above table, the logic value Rk denotes, as a function of itsvalue, the four sectors corresponding to the spans of values of μk. Itis therefore seen in this array that linear expressions can be obtainedbetween the coefficients h1, h2, h3 and μk. It is therefore possible tocalculate on line the coefficients of the interpolation filter on thebasis of μk without it being necessary to calculate them beforehand forany value of μ_(k) and to store them in a memory. This consequentlyresults in a considerable space saving.

Moreover, with the change of parameters (μ_(k) into μ′_(k)), a singlemultiplication by μ′_(k) may be necessary to calculate the contributionof a current input sample x_(k). Actually, by way of example, in thesector corresponding to R_(k)=0, the contribution h1·x_(k) is equal to(0.25−μ′_(k))x_(k), which may also be written 0.25·x_(k)−μ′_(k)x_(k). Itis therefore seen that a single multiplication by μ′_(k) may benecessary. Indeed, the multiplication of x_(k) by 0.25 is in fact adivision of x_(k) by 4, and this may very readily be achieved by a shiftof bits.

This therefore results in a significant reduction in the complexity ofimplementation and embodiment of the corresponding device. Thisembodiment therefore makes it possible here to avoid doing threemultiplications and to provide fast access to a memory. Furthermore, theonly multiplication performed is on a low number of bits.

An exemplary embodiment of a sampling frequency conversion device DCFwill now be described while referring more particularly to FIGS. 10 to13.

As illustrated in FIG. 10, the device DCF comprises a reception input MRable to receive the samples x_(k) of the digital signal, a formulatorMELB able to formulate the output samples y_(n), and an output SO1 todeliver these output samples y_(n).

The device DCF furthermore comprises a selector MSEL able to select thevalue of the ratio F_(s)/F_(i) from a suite of predetermined valuescorresponding to sampling frequency values compatible with severaldifferent standards, for example, the WiMAX or WiFi standards. It shouldbe noted here that the selector MSEL, in fact, selects the inverse ofthis ratio.

The formulator MELB includes a calculator MCL able to calculate the Ncontributions of each current input sample x_(k), and a processor ableto perform the updating of the values of the output samples.

In the exemplary embodiment described in FIG. 10, which corresponds to aparticular embodiment implementing the on-line calculation of thecoefficients of the interpolation filter on the basis of the parameterμ′_(k) in accordance with the above table, the calculator MCL includes adigitally controlled oscillator NCO that determines the temporalposition of a current input sample with respect to the N temporalinstants of interpolation respectively associated with the N outputsamples to be updated. The calculator also determines the values of theN coefficients of the filter on the basis of this temporal position. Inthe example described here, the temporal position is determined by wayof the parameter μ′_(k). Also, the first detector comprises a digitallycontrolled oscillator NCO capable, on the basis of the periods T_(s) andT_(i), and of a clock signal CK of period T_(s), delivered by a quartzQZT for example, of determining the parameter μ′_(k) for a current inputsample x_(k).

The calculator MCL also included a multiplier ML receiving the currentsample x_(k) and the parameter μ′_(k). The output of the multiplier aswell as the sample x_(k) are connected to two inputs of a block BLCcomprising adders ADD, subtractors SB, and a shifter for shifting bitsSH, all of which are individually selectable as a function of the valueof the temporal interval μ_(k), in other words, as a function of thelogic parameter R_(k) delivered by the oscillator NCO. In fact, in thisexample, the calculation of the values of the coefficients of the filterand the calculation of the contributions are done globally within theblock BLC.

The processor MT includes three inputs (N=3) IN1, IN2, IN3 to receive,respectively, the three contributions of a current input sample. Theprocessor can moreover be three adders ADD1, ADD2, ADD3, each possessinga first input EA1 coupled to the corresponding input of the processor(IN1, IN2 or IN3) and a second input EA2. The processor also includesthree multiplexers MUX1, MUX2, MUX3 to be controlled by controller MCfunctionally incorporated within the oscillator NCO, and each possessinga first input EMX1 connected to the output SA of the correspondingadder, and a second input EMX2.

The processor also includes a chain of three registers R1, R2, R3 eachpossessing an input ER1 coupled to the output of the correspondingmultiplexer and an output SR1 coupled to the second input EA2 of thecorresponding adder.

Moreover, the output of the last register of the chain, in this instancethe register R1, is coupled to the output SO1, while the output of eachof the other registers R2, R3 is coupled to the second input of themultiplexer associated with the following register in the chain ofregisters. More precisely, the output of the register R3 is coupled tothe second input of the multiplexer MUX2, while the output of theregister R2 is coupled to the second input of the register MUX1. Theprocessor also includes an auxiliary register RGA containing a zerovalue data item and whose output is coupled to the second input of themultiplexer MUX 3 associated with the first register R3 of the chain.

Reference is now made more particularly to FIGS. 11-13 to illustrate anexemplary manner of operation of the device of FIG. 10. The device DCFis regulated in time with the clock signal CK, an input sample x_(k)arriving, for example, at each rising edge of the clock signal CK.During a period of the clock signal CK, the three contributions of thecurrent input sample x_(k) are calculated and transmitted to theprocessor. The control signal DS which controls the multiplexers of theprocessor takes the value 1, thereby linking the first input of themultiplexers to their output, and therefore looping back the output ofeach of the registers to the second input of the corresponding adder.This makes it possible to accumulate in each of the registers thecorresponding contribution and to update the corresponding value of theoutput sample (FIG. 12).

Once this update has been performed, the oscillator NCO determines, byapplying the condition CD1 mentioned above, whether the sample y_(n)should or should not be delivered as output. If such is not the case, alogic signal DV takes the value “0”, indicating that the output of theregister R1 is not a valid data item. In the converse case, the signalDV rises to “1”, indicating that the output of the register R1 is valid.

In the example which is described here, output samples of rank n to n+2are updated and, in this regard, the three registers R1, R2 and R3, ineach of which the various contributions arising from the various inputsamples are accumulated for the output sample considered, are assignedrespectively to these three output samples.

When the sample y_(n) should be delivered, the value contained in theregister R1 is delivered and the registers R1 and R2 are assigned to thesamples y_(n+1) and y_(n+2), while the output sample y_(n+3) is assignedto the register R3. Also, the register R3 is initialized to a choseninitialization value, in this instance the value “0”, for example. Thisshift of registers is performed by controlling the signal DS, whichtakes the value “0”, for example, and which makes it possible to linkthe second inputs of the multiplexers to their corresponding outputs(FIG. 13). Of course, as long as the signal DV is at “0”, theconfiguration of FIG. 12 is maintained, accumulating the variouscontributions arising from the various input samples.

Not all the output samples receive an identical number of contributionsfor their updating. For example, as illustrated in FIG. 11, the edges ofthe signal DV have been slightly shifted with respect to the rising edgeof the clock signal CK. In the example described, the sample y₄ (on theI pathway and on the Q pathway) illustratively is ready to be deliveredon the first rising edge of the clock signal CK, which results in therise of the signal DV to “1”.

During the following cycle of the clock signal CK, the sample y₅ isupdated and is also ready to be delivered. Consequently, the signal DVstays at “1” and the sample y₅ is delivered on the following rising edgeof the clock signal CK.

In the course of the following cycle, the sample y₆ is further updatedand is also ready to be delivered on the following rising edge of theclock signal CK. The signal DV is still at “1”.

In the course of the following cycle CYC7, the following output sampley₇ is still being formulated. At the end of this cycle CYC7, the sampley₇ is not ready to be delivered since its formulation allows receivingthe contribution of a following input sample. The signal DV thereforefalls back to “0” and stays at this value during the following cycleCYC8, in the course of which the formulation of the output sample y₇continues by updating its value with two other contributions originatingrespectively from the following two input samples. It is only at the endof the cycle CYC8 that the sample y₇ is ready to be delivered, whichoccurs on the rising edge FM8 of the clock signal CK. The signal DV thenreverts to the value “1”.

In the course of the following cycle CYC9, the sample y₈ is delivered.The sample y₉ is updated in the course of the cycle CYC9, but thissample will require two more clock cycles of the signal CK to be updatedbefore being delivered. Then the sample y₁₀ and the sample y₁₁ isgenerally respectively one and two more clock cycles of the signal CK tobe updated before being delivered.

It is therefore seen that the output samples y_(n) are delivered in timewith the rising edges of the clock signal CK, but only on some of theserising edges as a function of the number of contributions for each ofthe output samples y_(n).

The number of contributions is different for the output samples. Eachnumber of contributions depends on N, on the conversion ratio, and alsoon μ_(k) (see condition CD1 above). In general the number ofcontributions is greater than or equal to N, and in practice, muchgreater than N.

The frequency of delivery of the output samples is therefore differentfrom the sampling frequency F_(i). From the signal processing viewpoint,the output samples y_(n) are representative of a sampling of the digitalsignal at a frequency F_(i).

Simulations show that the power consumed of a frequency conversiondevice in accordance with aspects varies substantially linearly with thefirst sampling frequency F_(s). On the other hand, for a given frequencyF_(s), this power consumed is substantially constant whatever the valueof the conversion ratio F_(s)/F_(i), whereas in the prior art approachesproviding for storage of filter coefficients for each conversion ratioenvisaged, the power consumed increases as the value of the conversionratio increases.

These simulation results are illustrated more particularly in FIGS. 14and 15 which relate to a device of the type of that of FIG. 10, embodiedwith 65 nm technology. In FIG. 14, curve C4 illustrates thesubstantially linear trend of the power as a function of F_(s). In FIG.15, curve C5 illustrates the substantially constant character of thepower as a function of the conversion ratio, for a frequency F_(s) equalto 60 MHz, while curve C6 is plotted for a frequency F_(s) equal to 120MHz. A substantially constant power of the order of 8 mW would beobtained for a frequency F_(s) equal to 240 MHz.

1. A method for converting a sampling frequency of a digital signalsampled at a first sampling frequency, the method comprising: receivingdigital signal input samples; and generating output samplescorresponding to a second sampling frequency based on the digital signalinput samples and an interpolation filter, the first sampling frequencybeing larger than the second sampling frequency; wherein generating theoutput samples comprises, for each of the digital signal input samples,updating current values of N successive output samples with Ncontributions, the N contributions being respectively calculated basedon a value of a current input sample of the digital signal input samplesweighted by values of N filter coefficients of the interpolation filterassociated with the current input sample, with N being an integergreater than 1, fixed and identical for all the digital signal inputsamples regardless of a value of a conversion ratio between the firstand second sampling frequencies.
 2. The method according to claim 1,wherein one of the output samples comprises a value updated with thecontribution originating from the current input sample when its valuehas been updated with a contribution from a current input sample and theoutput sample not receiving any contribution originating from afollowing input sample.
 3. The method according to claim 2, whereinupdating the current values of the N successive output samples, havingranks n to n+N−1 with n being an integer, comprises respectivelyassigning the N successive samples to N storage having ranks r₁ to r_(n)each N storage accumulating, for a considered output sample, thecontribution arising from the digital input samples, and wherein theoutput samples of a rank n comprise a value in the storage of rank r₁,assigning to registers of ranks r₁ to r_(n−l) to output samples of ranksn+1 to n+N−1, and initializing the register of rank r_(n) to a choseninitialization value.
 4. The method according to claim 1, wherein thedigital signal input samples are received and the output samples areformed in time with edges of a clock signal having a frequency equal tothe first sampling frequency, and the output samples are delivered on aportion of the edges of the clock signal.
 5. The method according toclaim 1, wherein the value of the conversion ratio is selected fromnon-integer values.
 6. The method according to claim 1, wherein thevalue of the conversion ratio is selected from values corresponding tovalues of second sampling frequencies that are compatible with at leastone of a WiMax and WiFi standard.
 7. The method according to claim 1,wherein a width of an impulse response of the interpolation filter fixesa value N.
 8. The method according to claim 1, wherein the interpolationfilter has an impulse response width equal to the product of amultiplying coefficient, and a second time period corresponding to thesecond sampling frequency, and wherein a value of N is fixed at a valueequal to a higher integer part of the multiplying coefficient.
 9. Themethod according to claim 1, wherein the interpolation filter has anamplitude/frequency response curve having zeros at frequencies which aremultiples of the second sampling frequency.
 10. The method according toclaim 9, wherein the amplitude/frequency response curve is a form of aproduct of a cardinal sine function squared times a cosine function. 11.The method according to claim 1, wherein the calculation of the Ncontributions of each current input sample comprises determining atemporal position of the current input sample with respect to N temporalinstants of interpolation, spaced mutually apart by a second time periodcorresponding to the second sampling frequency and respectivelyassociated with the N updated successive output samples, and thedetermination of the values of the N filter coefficients being based onthe temporal position.
 12. The method according to claim 11, whereineach of the digital signal input samples arrive at a first temporalinstant of interpolation, and determining the temporal position of acurrent input sample with respect to the N temporal instants ofinterpolation comprises determining a temporal interval between thefirst temporal instant associated with the current input sample and oneof the N temporal instants of interpolation, and wherein determiningother temporal intervals between the first instants and N−1 othertemporal instants of interpolation is based on the temporal interval,and the values of the N coefficients of the interpolation filtercorrespond respectively to values of an impulse response of theinterpolation filter at the temporal interval.
 13. The method accordingto claim 12, wherein determining each of the N filter coefficients ofthe interpolation filter is based on a function of the temporalinterval.
 14. The method according to claim 13, wherein the functioncomprises at least one linear zone.
 15. The method according to claim14, wherein a value of the temporal interval varies between 0 and 1,wherein an auxiliary temporal interval is based on a temporal intervalhaving a value equal to a span [0; 0.25], and wherein calculating acontribution of a current input sample comprises a single multiplying ofthe value of the current input sample with a value of the auxiliarytemporal interval.
 16. A method for processing an analog signal,comprising: performing an analog/digital conversion of the analogsignal; receiving digital signal input samples from the analog/digitalconversion; and generating output samples corresponding to a secondsampling frequency based on the digital signal input samples and aninterpolation filter, the first sampling frequency being larger than thesecond sampling frequency; wherein generating the output samplescomprises, for each of the digital signal input samples, updatingcurrent values of N successive output samples with N contributions, theN contributions being respectively calculated based on a value of acurrent input sample of the digital signal input samples weighted byvalues of N filter coefficients of the interpolation filter associatedwith the current input sample, with N being an integer greater than 1,fixed and identical for all the digital signal input samples regardlessof a value of a conversion ratio between the first and second samplingfrequencies.
 17. The method according to claim 16, wherein theanalog/digital conversion comprises a delta-sigma type conversion. 18.The method according to claim 16, wherein the analog signal is selectedto be compatible with at least one of a WiMAX and WiFi standard.
 19. Amethod for converting a sampling frequency of a digital signal sampledat a first sampling frequency, the method comprising: receiving digitalsignal input samples; and generating output samples corresponding to asecond sampling frequency based on the digital signal input samples withthe first sampling frequency being larger than the second samplingfrequency; wherein generating the output samples comprises updatingcurrent values of N successive output samples with N contributions, withN being an integer greater than 1, the N contributions beingrespectively calculated based on a value of a current input sample ofthe digital signal input samples weighted by values of N filtercoefficients associated with the current input sample.
 20. The methodaccording to claim 19, wherein one of the output samples comprises avalue updated with the contribution originating from the current inputsample when its value has been updated with a contribution from acurrent input sample and the output sample not receiving anycontribution originating from a following input sample.
 21. The methodaccording to claim 20, wherein updating the current values of the Nsuccessive output samples having ranks n to n+N−1, with n being aninteger, comprises respectively assigning the N successive samples to Nstorage having ranks r₁ to r_(n), each N storage accumulating, for aconsidered output sample, the contribution arising from the digitalinput samples and wherein the output samples of rank n comprise a valuein the storage of rank r₁, assigning to registers of ranks r₁ to r_(n−1)to output samples of ranks n+1 to n+N−1, and initializing the registerof rank r_(n) to a chosen initialization value.
 22. The method accordingto claim 19, wherein the digital signal input samples are received andthe output samples are formed in time with edges of a clock signalhaving a frequency equal to the first sampling frequency, and the outputsamples are delivered on a portion of the edges of the clock signal. 23.A device for converting the sampling frequency of a digital signalsampled at a first sampling frequency, the device comprising: a receiverfor receiving digital signal input samples; a generator for generatingthe output samples corresponding to a second sampling frequency based onthe digital signal input samples and an interpolation filter, the firstsampling frequency being larger than the second sampling frequency; andthe generator comprising a calculator for calculating N contributionsfor a current digital signal input sample of the digital signal inputsamples respectively based on a value of the current digital signalinput sample weighted by values of N filter coefficients associated withthe current input sample, with N being an integer greater than 1, fixedand identical for all the digital signal input samples regardless of avalue of a conversion ratio between the two frequencies, and a processorfor updating current values of N successive output samples with the Ncontributions for each digital signal input sample.
 24. The deviceaccording to claim 23, wherein the generator further comprises acontroller for authorizing the delivery of an output sample having avalue updated with a contribution originating from the current inputsample when its value has been updated with a contribution from acurrent input sample and an output sample not receiving any contributionoriginating from a following input sample.
 25. The device according toclaim 24, wherein the processor comprises: N inputs for receivingrespectively N contributions of a current input sample; N adders eachhaving a first input coupled to the corresponding input of the processorand a second input; N multiplexers being controlled by the controllerand each having a first input coupled to an output of a correspondingadder and a second input; a chain of N registers each possessing aninput coupled to the output of a corresponding multiplexer and an outputcoupled to the second input of the corresponding adder; the output of alast register of the chain of N registers being coupled to the output ofeach other register, each other register being coupled to the secondinput of the multiplexer associated with a following register in thechain of N registers; and an auxiliary register including a zero valuedata item and coupled to the second input of a multiplexer associatedwith a first register of the chain of N registers.
 26. The deviceaccording to claim 23, wherein the receiver is for receiving the digitalsignal input samples in time with edges of a clock signal having afrequency equal to the first sampling frequency; wherein the generatoris for generating the digital signal output samples in time with theedges of the clock signal; and wherein the digital signal output samplesare delivered on a portion of the edges of the clock signal.
 27. Thedevice according to claim 23, further comprising a selector forselecting the value of the conversion ratio from non-integer values. 28.The device according to claim 23, further comprising a selector forselecting the value of the conversion ratio from values of secondsampling frequencies that are compatible with at least one of the WiMaxand WiFi standard.
 29. The device according to claim 23, wherein a widthof an impulse response of the interpolation filter fixes a value of N.30. The device according to claim 23, wherein the interpolation filterhas an impulse response width equal to the product of a multiplyingcoefficient, and a second time period corresponding to the secondsampling frequency, and wherein a value of N is fixed at a value equalto a higher integer part of the multiplying coefficient.
 31. The deviceaccording to claim 23, wherein the interpolation filter has anamplitude/frequency response curve having zeros at frequencies which aremultiples of the second sampling frequency.
 32. The device according toclaim 31, wherein the amplitude/frequency response curve is a form of aproduct of a cardinal sine function squared times a cosine function. 33.The device according to claim 23, wherein the calculator comprises afirst detector for determining a temporal position of the currentdigital input sample with respect to N temporal instants ofinterpolation, respectively associated with the N successive outputsamples to be updated, and a second detector for determining the valuesof the N filter coefficients based on the temporal position.
 34. Thedevice according to claim 33, wherein the first detector determines atemporal interval between a first instant associated with the currentdigital signal input sample and one of the N temporal instants ofinterpolation, for determining other temporal intervals between thefirst instant and N−1 other temporal instants of interpolation is basedon the temporal interval; and wherein the values of the N coefficientsof the interpolation filter correspond respectively to values of animpulse response of the interpolation filter at the temporal intervals.35. The device according to claim 34, wherein the second detectordetermines each of the N filter coefficients of the interpolation filterbased on a function of the temporal interval.
 36. The device accordingto claim 35, wherein the function of the temporal interval comprises atleast one linear zone.
 37. The device according to claim 36, wherein avalue of the temporal interval varies between 0 and 1 and the firstdetector is for determining an auxiliary temporal interval based on atemporal interval having a value equal to [0; 0.25], and wherein thecalculator comprises a single multiplier having an input coupled to thereceiver and a second input coupled to the first detector for receivingthe auxiliary temporal interval.
 38. The device according to claim 37,wherein the calculator further comprises a block comprising at least oneadditional adder, at least one subtractor and at least one shifter forshifting bits individually selectable as a function of the value of thetemporal interval.
 39. A device for processing an analog signalcomprising: an input for receiving the analog signal; an analog/digitalconverter coupled to said input and having an output for outputting adigital signal; and a sampling frequency conversion device having asampling frequency conversion device input that is coupled to theanalog/digital converter output and comprising a receiver for receivinginput samples of the digital signal, an interpolation filter, and agenerator for generating the output samples corresponding to a secondsampling frequency based upon the input samples and of the interpolationfilter, the first sampling frequency being larger than the secondsampling frequency, the generator comprising a calculator forcalculating N contributions for each current input sample respectivelybased on a value of the current input sample weighted by values of Nfilter coefficients of the interpolation filter associated with thisinput sample, with N being an integer greater than 1, fixed andidentical for all the input samples whatever the value of a conversionratio between the two frequencies, and a processor for updating thecurrent values of N successive output samples with the N contributionsfor each input sample.
 40. The device according to claim 39, wherein theanalog/digital converter comprises a delta-sigma type analog/digitalconverter.
 41. A wireless communications system receiver comprising: anantenna for receiving an incident signal; and an analog/digitalconverter coupled to said antenna; and a sampling frequency conversiondevice coupled to the analog/digital converter and comprising a receiverfor receiving input samples of the incident signal, an interpolationfilter, and a generator for generating output samples corresponding to asecond sampling frequency based upon the input samples and theinterpolation filter, the first sampling frequency being larger than thesecond sampling frequency, and the generator comprising a calculator forcalculating N contributions for each current input sample respectivelybased on a value of the current input sample weighted by values of Nfilter coefficients of the interpolation filter associated with thisinput sample, with N being an integer greater than 1, fixed andidentical for all the input samples whatever the value of a conversionratio between the two frequencies, and a processor for updating currentvalues of N successive output samples with the N contributions for eachinput sample.
 42. The wireless communications system receiver accordingto claim 41, wherein said antenna is for receiving multi-standardincident signals.
 43. The wireless communications system receiveraccording to claim 42, wherein one of the multi-standard incidentsignals is selected to be compatible with at least one of a WiMAX andWiFi standard.